Circuits and methods to linearize conversion gain in a dc-dc converter

ABSTRACT

Described examples include DC-DC power conversion systems, apparatus and methods for linearizing a DC-DC circuit conversion gain, including a gain circuit providing an output signal according to a gain value and the difference between a first compensation signal and a threshold signal, and a switching circuit selectively operative when the first compensation signal exceeds the threshold signal to linearize the conversion gain by providing a second compensation signal for pulse width modulation of at least one DC-DC converter switch according to the threshold signal and the gain circuit output signal.

CROSS REFERENCE TO RELATED APPLICATIONS

Under 35 U.S.C. § 120, this continuation application claims benefits ofand priority to U.S. patent application Ser. No. 14/720,641 (TI-75028),filed on May 22, 2015, which claims priority to and the benefit of U.S.Provisional Patent Application Ser. No. 62/002,475, filed May 23, 2014.The entirety of the above referenced applications is hereby incorporatedherein by reference.

TECHNICAL FIELD

The present disclosure relates to power conversion and more particularlyto circuits and methods to linearize conversion gain in a DC-DCconverter.

BACKGROUND

CMOS technology is being aggressively scaled to reduce physicaldimensions and supply voltage to meet low power, low area and highperformance specifications in portable electronics and otherapplications. Also, new battery chemistries seek to extend device usageto lower voltages. However, certain loads like piezo-electric speakers,LED drivers, micro electromechanical device (MEMs) sensors, Cameraflash, USB on-the-go (USB-OTG) circuitry, etc. require regulated highvoltages for proper operation, and are generally powered using boostconverters or buck-boost converters. Reduced system and battery voltageswhich serve as input supplies result in higher conversion ratios forthese converters. Certain system features employing dynamic voltagescaling (DVS) also result in variable conversion ratios. There is agrowing demand for high conversion gain boost and buck-boost converters.The conversion gain M of a boost converter is given as the ratio of theoutput voltage VOUT to the input voltage VIN (M=VOUT/VIN=1/(1−D)). D isthe duty cycle or duty ratio representing the percentage of time thatthe main DC-DC converter switch is on, and higher conversion gain Mresults from higher duty cycle operation. The converter small signalgain G_(C) depends upon the duty ratio D (G_(C)=δVOUT/δD=1/(1−D)²),where the gain G_(C) is non-linear and increases with rising dutycycles. At high converter gain, small variations in the duty cycleresults in large converter output voltage changes and increasedlikelihood of converter instability. Higher duty cycle operation alsoresults in increased di/dt, increased EMI and higher output voltagesensitivity to duty cycle changes due to noise and jitter. High gainstability and noise issues have previously been addressed by feedbacklinearization or pre-distortion. Feedback linearization requires compleximplementations and high digital hardware cost. Many pre-distortionsystems employ low pass filters which increase the loop delay, as wellas inverse computation in the analog domain which is typicallyinaccurate. Some pre-distortion techniques also use analog multipliersin the feedback loop which increases circuit cost and space. Anotherapproach uses modulated ramps, and works well for open loop controlwhere output voltage accuracy is not a concern. However, use ofmodulated ramp techniques in closed loop application requires controlcurrent generation using an op-amp or inductor for modulating the ramp,and this approach is not well suited for hysteretic mode control whereswitching period is variable with load and hence affects control gainG_(C).

SUMMARY

In described examples, DC-DC power conversion systems and linearizationapparatus include a gain circuit providing an output signal according toa gain value and the difference between a first compensation signal anda threshold signal. In certain examples, the gain circuit output signalis based on a ratio of the difference between the first compensationsignal and the threshold signal divided by the gain value. In oneexample, the gain value is greater than unity and is adjustable orprogrammable. A switching circuit selectively operates to linearize thesystem conversion gain by providing a second compensation signal forpulse width modulation of at least one DC-DC converter switch accordingto the threshold signal and the gain circuit output signal when thefirst compensation signal exceeds the threshold signal. When the firstcompensation signal is below the threshold signal, the secondcompensation signal is provided according to the first compensationsignal. Certain examples include a circuit to selectively adjust thethreshold signal. In other described examples, a method of linearizing aDC-DC converter gain includes receiving a first compensation signal,operating the DC-DC converter according to a second compensation signal,generating a gain circuit output signal according to a non-zero gainvalue and a difference between the first compensation signal and athreshold signal. The method further includes providing the secondcompensation signal according to the first compensation signal if thefirst compensation signal is less than the threshold signal, andselectively providing the second compensation signal according to thethreshold signal and the gain circuit output signal if the firstcompensation signal is greater than the threshold signal.

DESCRIPTION OF THE VIEWS OF THE DRAWINGS

FIG. 1 is a schematic diagram of a DC-DC power conversion systemincluding a boost converter stage and a linearization apparatus in lowduty cycle operation in a first mode for pulse width modulation of atleast one DC-DC converter switch according to a first compensationsignal representing a difference between a DC-DC converter output and areference signal.

FIG. 2 is a schematic diagram of the system with the linearizationapparatus operating in a second mode for higher duty cycles toselectively linearize the conversion gain by pulse with modulationaccording to a threshold signal and a gain circuit output signal.

FIG. 3 is a graph of conversion gain as a function of the firstcompensation signal for various gain values in the system of FIGS. 1 and2.

FIG. 4 is a graph of conversion gain as a function of duty cycle for aunity gain value in the system of FIGS. 1 and 2.

FIG. 5 is a graph of small signal gain as a function of duty cycle ratioand the first compensation signal for a unity gain value in the systemof FIGS. 1 and 2.

FIG. 6 is a schematic diagram of another power conversion system exampleincluding a flyback DC-DC converter stage and a linearization apparatusfor selectively linearizing the DC-DC conversion gain.

FIG. 7 is a graph of conversion gain as a function of the firstcompensation signal for various gain values in a buck-boost conversionsystem.

FIG. 8 is a schematic diagram of a DC-DC power conversion system with alinearization apparatus using digital control.

FIG. 9 is a schematic diagram of an adjustable threshold circuit forselectively adjusting the threshold signal in the systems of FIGS. 1, 2and 6.

FIG. 10 is a schematic diagram of an adjustable gain circuit forselectively adjusting the gain value in the system of FIGS. 1, 2 and 6.

FIG. 11 is a flow diagram of a method of linearizing a DC-DC conversiongain.

DETAILED DESCRIPTION

In the drawings, like reference numerals refer to like elementsthroughout, and the various features are not necessarily drawn to scale.

FIGS. 1 and 2 show a power conversion system 100 with a boost DC-DCconverter stage or circuit 101 having an input receiving a DC inputsignal from a DC input source or supply 102, such as a DC input voltagesignal VIN. The circuit 101 also includes a converter output providing aDC output signal (e.g., voltage) VOUT to power a connected load R. TheDC-DC converter circuit 101 is a boost converter with an inductor Lcoupled between the input and a switching node, with a switch S4connected between the switching node and a constant voltage node GND,and a second switch S5 connected between the switching node and apositive output node. In other examples, the switch S5 can be replacedwith a diode (not shown) to implement a boost converter circuit 101 withat least one switching device (S4) coupled between the input and outputof the converter circuit 101. An output capacitance C is connectedbetween the positive output terminal and GND. Examples of suitableswitching devices S4 and S5 and other switches in the system 100 includebipolar transistors, MOSFETs, IGBTs, IGCTs and other devices operableaccording to a corresponding switching control signal to be in a firstconductive state or a second non-conductive state. The switches S4 andS5 of the boost converter circuit 101 are driven by switching controlsignals d and d′ provided by a driver and dead-time logic circuit 104along switching control lines 106 and 108, respectively. The switchingcontrol signals d and d′ are provided from the circuit 104 as pulsewidth modulated (PWM) signals for selectively opening and closing thecorresponding switches S4 and S5 according to a pulse width modulatedsignal “PWM” provided from a linearization circuit 110 along aconnection 122. The circuit 104 provides the signals d and d′ incomplimentary fashion to selectively turn the main boost converterswitch S4 on while S5 is turned off to store energy in the inductor L,and S4 is then turned off while S5 is turned on to charge the outputcapacitance C using current from the inductor L. In one example, thecircuitry 104 provides a controlled “dead-time” between switching statechanges during which both switches S4 and S5 are off. Pulse widthmodulated switching operation of S4 and S5 (or just S4 with S5 replacedby a diode) controls the amplitude of the DC output signal VOUT.

The output voltage VOUT is sensed by a feedback circuit including aresistive voltage divider formed by resistors R2 and R3 connectedbetween the positive DC output node and GND to provide a feedbackvoltage signal VFB. The feedback signal VFB is provided to acompensation circuit including an op amp 112 with an input impedance Z1connecting the feedback voltage signal VFB to an inverting input (−) anda feedback impedance Z2 connected between the inverting input and the opamp output to provide a first compensation signal, in this example avoltage signal Vc to the linearization circuit 110. The non-inverting(+) input of the op amp 112 is provided with a reference voltage signalVREF from a voltage source 114. The feedback voltage divider circuit R2,R3 and the compensation circuit impedances Z1 and Z2 set the amplitudeof the first compensation signal Vc according to the DC output signalVOUT and the reference signal VREF. The reference signal VREF operatesas a setpoint for closed loop regulation of the converter output voltagesignal VOUT. The first compensation signal Vc at any given timerepresents an output error or difference between the reference signalVREF and the DC-DC converter output signal VOUT scaled by the resistivedivider as the feedback voltage signal VFB.

The linearization apparatus 110 includes a gain circuit 150 with inputsreceiving the first compensation signal Vc and a threshold signal, suchas a voltage Vcq from a threshold circuit 140. Unless otherwisespecified herein, a signal can be an analog current or voltage, or adigital value. The gain circuit 150 includes an output 152 whichprovides a gain circuit output signal, such as a current signal I1 (FIG.2), according to a non-zero gain value G_(A) and the difference Vc−Vcqbetween the first compensation signal Vc and the threshold signal Vcq.The linearization circuit 110 selectively uses the gain circuit outputsignal I1 at high duty cycle operation to at least partially linearizethe DC-DC converter circuit conversion gain M representing theoutput-to-input ratio of the system 100 (e.g., M=VOUT/VIN). At firstcompensation signal levels Vc less than the threshold Vcq, thelinearization circuit 110 operates in a first mode to provide the signalPWM to the driver and dead-time logic circuit 104 according to the firstcompensation signal Vc. At higher values of Vc greater than or equal tothe threshold signal Vcq, the linearization circuit 110 provides thesignal PWM according to the threshold signal Vcq and the gain circuitoutput signal I1 in a second operating mode. In the example of FIGS. 1and 2, the closed loop control of the system 100 is therefore adjustedfor higher duty cycle operation to linearize the gain M by adapting theduty cycle D as a function of the gain value G_(A) and the differenceVc−Vcq. In one example, the pulse width modulation is controlled in thesecond mode according to a second compensation signal Vca which isproportional to (Vc−Vcq)/G_(A). The gain value G_(A) is adjustable insome implementations, for example as shown in FIGS. 8 and 10 below. Inone example, the gain value G_(A) is greater than 1.0. In this manner,the overall system conversion gain M is selectively lowered for dutycycles D above a threshold value D_(Q) where D_(Q) corresponds to Vcq.

In the example of FIGS. 1 and 2, the linearization apparatus 110implements dual mode operation for selectively adapting the system gainfor higher duty cycles using a switching circuit including switchesS1-S3 as well as PMOS transistors MP5 and MP6, and a PWM circuitincluding a comparator 120, a resistor R1 in a ramp generator circuit128. The comparator 120 includes a non-inverting (+) first input 124receiving the second compensation signal Vca established by a current I4flowing from the first input node 124 to GND through the resistor R1.The switches S1-S3 selectively provide one or more currents to thesumming junction formed at the node 124 according to the operating modeof the linearization apparatus 110 to establish the second compensationsignal Vca as a voltage at the node 124. The ramp generator 128 providesa ramp or other repeating voltage signal to an inverting (−) secondinput 126 of the comparator 120. The comparator 120 includes an output122 operative to provide the signal PWM to control the output signalVOUT of the DC-DC converter circuit 101. The voltage across theresistance R1 provides the second compensation signal Vca as a voltageat the first comparator input 124 for comparison with the ramp signal atthe second comparator input 126. The continuous comparison of thevoltages at the inputs 124 and 126 provides a pulse width modulatedsignal PWM from the comparator output to the driver and dead-time logiccircuit 104 to generate the switching control signals d and d′ on thelines 106 and 108 for operating the boost converter switches S4 and S5.

The linearization circuit apparatus 110 also includes a secondcomparator circuit 130 receiving the first compensation signal Vc andthe threshold signal Vcq at + and − inputs, respectively. The comparator130 has an output 132 providing a mode control signal to the switches S1and S3, and also to an inverter 134 to operate the switch S2 to controlthe operating mode of the linearization apparatus 110. The comparator130 places the linearization apparatus 110 in the first mode if Vc isbelow Vcq, and places the linearization apparatus 110 in the second modeif Vc is greater than or equal to Vcq.

The switching circuit includes a first switch S1 operative in the firstmode disconnect the output 152 of the gain circuit 150 from the firstcomparator input 124, and in the second mode to provide a first currentsignal I1 from the gain circuit 150 to the first comparator input 124. Asecond switch S2 allows a second current I2 to flow into the firstcomparator input 124 in the first mode. The second current I2 isproportional to the first compensation signal Vc by operation of thetransistor MP5 with a source connected through the switch S2 to apositive supply voltage V+ at a positive voltage node 118, and a drainconnected to the non-inverting first comparator input 124. The gate ofMP5 is driven by the first compensation signal voltage Vc and thus thecurrent I2 flowing into the comparator input 124 in the first mode isproportional to the first compensation signal Vc. In the second mode,the second switch S2 is open or off to prevent flow of the secondcurrent I2 into the first comparator input 124. The switching circuitalso includes a third switch S3 connected in series with the transistorMP6 between V+ and the comparator input 124. The transistor MP6 has agate controlled by the threshold signal voltage Vcq, and when the switchS3 is closed (FIG. 2), a corresponding current I3, which is proportionalto the threshold signal Vcq, flows into the resistor R1 along with thefirst current I1 from the gain circuit 150 (through capital S1).

The gain circuit 150 has an output 152 providing the gain circuit outputsignal as a current I1 in this example based on a ratio of thedifference Vc−Vcq between the first compensation signal Vc and thethreshold signal Vcq to the gain value G_(A). The gain circuit 150includes a transconductance amplifier (GMA) 116 receiving the firstcompensation signal Vc and the threshold signal Vcq at + and − inputs,respectively, along with a differential current mirror circuit formed byN-channel transistors MN1 and MN2 and P-channel transistors MP1, MP2,MP3 and MP4. The relative sizing of the current mirror transistors andthe gain circuit 150 provides for scaling of the transconductanceamplifier output currents IOUT+ and IOUT− by the gain value G_(A) suchthat the resulting gain circuit output signal current I1 is proportionalto (Vc−Vcq)/G_(A). In one example, the transistors MP1, MP2 and MN1 forma unity gain current mirror with a current flowing through MP2 and inMN1 equal to IOUT− from the transconductance amplifier 116. Thetransistors MP1, MP2 and MN1 in one example are individually fabricatedusing an integer number N transistors substantially equal in terms ofwidth w and length l. The transistor MP3 is also of the same size(N*w/l). Transistors MP4 and MN2 are smaller in one example,individually formed using an integer number N/G_(A) transistors of sizew/l. The transistors MN1 and MN2 form a current mirror to conduct acurrent through MN2 proportional to the current through MN1 scaled bythe gain value G_(A) (e.g., IOUT−/G_(A)). MP3 and MP4 form a currentmirror with respect to the transconductance amplifier output currentIOUT+ flowing in MP3, and the current flowing through MP4 isproportional to the current through MN3 scaled by the gain value G_(A)(e.g., IOUT+/G_(A)). This establishes a differential current signal I1at the gain circuit output node 152 joining MP4 and MN2 which isproportional to (Vc−Vcq)/G_(A). During high duty cycle operation(D>D_(Q), Vc>Vcq), the comparator 130 enables switches S1 and S3 anddisables switch S2. The voltage to current converter or transconductanceamplifier 112 is used to convert the difference between Vc and Vcq tocurrent, and the current is reduced in the gain circuit 150 by the gainvalue G_(A) to obtain a linearized duty ratio D_(Q+)ΔD/G_(A).

Where G_(A) is greater than 1.0, the gain circuit 150 provides the gaincircuit output signal as the first current I1 generally proportional to(Vc−Vcq)/G_(A). In this example, the relative sizing of the transistorsin the gain circuit 150 sets the gain value G_(A). Different techniquescan be used in other examples to set a non-unity gain value, includingadjustable and/or programmable gain values, for example as shown in FIG.10 below. Operation of the comparator 130 and the switches S1-S3provides the current I2 through the resistor R1 such that the PWM signalprovided by the comparator 120 has a duty cycle set according to thefirst compensation signal Vc in the first mode when Vc is less than Vcq.FIG. 1 shows the switching circuit configuration during the first mode.In the second mode when Vc is greater than or equal to Vcq, S1 and S3are closed and S2 is open as shown in FIG. 2. The current I4 through theresistor R1 is the sum of the currents I1 and I3 in the second mode, andthe resulting second compensation signal voltage Vca at the firstcomparator input node 124 is proportional to Vcq+(Vc−Vcq)/G_(A).

FIGS. 3-5 illustrate graphs of conversion gain and small signal gain asa function of the first compensation signal Vc and the correspondingduty cycle ratio D at various operating points in the power conversionsystem 100 of FIGS. 1 and 2. A graph 300 in FIG. 3 shows the systemconversion gain M (M=VOUT/VIN) as a function of the first compensationsignal Vc for various gain values in the system of FIGS. 1 and 2. In theexample of FIGS. 3-5, the first compensation voltage Vc is establishedby the resistors R2 and R3 (FIGS. 1 and 2) where Vc=1.0 V represents100% duty cycle (D=1.0) for a unity gain value G_(A)=1.0. Also, thethreshold signal Vcq is set to a voltage of 0.5 V in this example,corresponding to 50% duty cycle ratio D for unity gain value G_(A)=1.0.Other voltage values and scaling can be used in other examples. Incertain examples, different threshold signal levels Vcq can be used, andthe threshold signal Vcq is adjustable and/or programmable in certainembodiments, for example, as shown in FIG. 9 below.

Curve 301 in FIG. 3 shows operation of the linearization apparatus 110in the first mode from 0.0≤Vc<0.5 where the system conversion gain Mfollows a rising conversion gain profile corresponding to a unity gainvalue G_(A)=1.0. Curve 302 shows continuation of this conversion gain Mfor Vc≥0.5 with G_(A)=1.0. A graph 400 in FIG. 4 shows a conversion gaincurve 401 as a function of duty cycle ratio for a unity gain value, andthe graph 500 in FIG. 5 illustrates a small signal gain curve 501 as afunction of duty cycle ratio D and the first compensation signal Vc fora unity gain value in the system 100 of FIGS. 1 and 2. As seen in thecurves 302, 401 and 501 in FIGS. 3-5, duty cycle values 0.5≤D≤0.8correspond to 2.0≤M≤5.0, with the conversion gain curve 401 risingsignificantly above Vc=0.7. At unity gain value G_(A), therefore, largeconversion gains M higher than 5 are largely impractical due to need forextreme duty cycle operation, and the steep rise in the small signalgain curve 501 demonstrates the susceptibility to instability and noiseat such high duty cycle operation.

To address these shortcomings, the linearization apparatus 110advantageously allows the gain value G_(A) to be set in certain examplesgreater than unity. This, in turn, extends the rising conversion gaincurves to higher first compensation signal levels Vc. For example, thecurve 303 in FIG. 3 shows system conversion gain with a gain valueG_(A)=2.0, curve 304 shows system conversion gain for a gain valueG_(A)=4.0, and curve 305 shows the conversion gain M using a gain valueG_(A)=5.0. This feature of the system 100 and the linearizationapparatus 110 facilitates improved stability and control for high dutycycle operation above the threshold signal level Vcq. Although thecomposite curves 301/303, 301/304 and 301/305 shown in FIG. 3 are notstrictly linear, these composite curves are referred to herein as“linearized” as the composite curves are significantly more linear withrespect to conversion gain M as a function of the first compensationsignal Vc) that is the composite unity gain value curve 301/302. Theselectively linearized system gain adaptation of the linearizationapparatus 110 reduces the output variations in response to small dutycycle changes at higher conversion gains to enhance system stability andmitigate electromagnetic interference, noise and jitter in the system100. At the same time, extension of the control range renders the system100 more easily adaptable for use with smaller input voltages VIN (e.g.,lower voltage battery sources, etc.) while providing output voltagesVOUT at relatively high amplitudes useful for driving piezo-electricspeakers, LED drivers, micro electromechanical device (MEMs) sensors,camera flash, USB on-the-go (USB-OTG) or other loads requiring voltagesabove the input supply level.

The apparatus 110 thus provides conversion gain linearization withselectively lowered gain M at higher duty cycle values D above athreshold D_(Q) corresponding to the threshold signal Vcq. In operation,the difference Vc−Vcq between the current operating point Vc and thethreshold Vcq is determined by the apparatus 110 and linearized with thechosen slope or gain value G_(A). The difference ΔD between D and D_(Q)is computed (ΔD=D−D_(Q)), which is used to compute a new linearized dutycycle D as Dq+(ΔD/G_(A)). The adjusted or linearized conversion gainM′=VOUT′/VIN′ is given by the following equation (1):

$\begin{matrix}{{M^{\prime} = {{\frac{V_{{out},Q}}{\left( {1 - D_{Q}} \right)}\frac{1}{{s^{2}{LC}} + {\frac{L}{R}s} + \left( {1 - D_{Q}} \right)^{2}}} + {\frac{\Delta \; V_{out}}{\left( {1 - \frac{\Delta \; D}{G_{A}}} \right)}\frac{1}{{s^{2}{LC}} + {\frac{L}{R}s} + \left( {1 - \frac{\Delta \; D}{G_{A}}} \right)^{2}}}}},} & (1)\end{matrix}$

where L is the value of the DC-DC converter inductor L (FIGS. 1 and 2),C is the converter output capacitance C and “S” is the Laplace transferoperator.

The control transfer function thus provides unmodified behavior untilD_(Q) (corresponding to Vcq) to retain the useful part of highconversion gain curve and to linearize above this threshold point. Thecurves 303-305 in FIG. 3 show the new linearized conversion gain M as afunction of the modified first compensation voltage V_(C) for gainvalues G_(A) of 2.0, 4.0 and 5.0. The modified conversion gain followsthe G_(A)=1.0 curve 301 until the Vcq threshold and then follows a morelinear path with a slope determined by G_(A) for higher compensationvoltages Vc. In one example, scaling of the gain circuit current mirrortransistors and the threshold signal setting Vcq allows the system 100to be tailored to provide a linearization profile suitable for a givenapplication.

The gain circuit 150 includes a current mirror circuit to provide thegain circuit output signal as a current signal I1 proportional to thedifference Vc−Vcq between the first compensation signal Vc and thethreshold signal Vcq. The current mirror circuit in some examplesincludes a plurality of selectable transistors to selectively adjust thegain value G_(A), for example in FIG. 10 below. The example gain circuit150 in the system 100 of FIGS. 1 and 2 uses a differentialtransconductance amplifier 116 and current mirror circuitry along withthe switching circuit to implement the adjusted second compensationsignal Vca based on the difference Vc−Vcq between the first compensationsignal Vc and the threshold signal Vcq with scaling by the gain valueG_(A). Different circuits can be used in other examples, whether basedon current signals or voltage signals or combinations of currents andvoltages. For example, summing amplifiers and difference amplifiers areused in other examples to implement the selective linearizationoperation with fixed or adjustable gain values G_(A) and/or fixed oradjustable thresholds Vcq.

Referring now to FIGS. 6 and 7, other embodiments are possible in whichthe linearization apparatus is used in combination with other forms ofDC-DC converter stage. FIG. 6 illustrates another power conversionsystem example 100 including a flyback converter stage circuit 601 witha switch S4 connected in series with a primary winding of a transformerTI between the input source 102 and circuit ground GND, as well as aninductor LM connected in parallel with the transformer primary. Theswitch S4 in this case is operated according to a duty cycle controlledsignal d from the driver on line 106. A secondary winding of thetransformer TI provides a rectified DC output across the outputcapacitance C via a rectifier diode D1 with an anode connected to thesecondary winding and a cathode connected to the positive outputterminal. The example in FIG. 6 also includes a linearization apparatus110 as described above. An alternate example can use activerectification to replace the rectifier diode D1 of FIG. 6 with a switchfor rectifying the output of the transformer secondary to provide arectified DC voltage output across the output capacitance C.

In another example, the disclosed linearization apparatus and techniquescan be used with a non-inverting buck-boost converter stage topology(buck cascaded with boost, not shown) when operating in boost mode.Synchronous non-inverting buck-boost examples are possible using a pairof switches, and other non-inverting buck-boost circuits can be used incombination with the described linearization circuitry 110 which canemploy any suitable combination of at least one DC-DC converter switchwith one or more diodes or further switches for DC-DC conversionoperable at unity or higher gain. FIG. 7 provides a graph 700 showingconversion gain M including curves 701-705 as a function of the firstcompensation signal Vc for various gain values 1.0, 2.0, 4.0 and 5.0 ina buck-boost conversion system example. In this example, Vcq is againset to 0.5 V and the duty cycle D is 100% at 1.0 V for a gain valueG_(A)=1.0. Unlike the boost converter example of FIGS. 1-5, thebuck-boost converter circuit allows initial operation at low duty cycles(e.g., 0.0≤Vc≤0.3) at conversion gain values M below 1.0 (e.g., “buck”mode) with the output voltage VOUT less than the input voltage VIN. Thisis shown in the curve portion 701 which initially begins below 1.0 atlow duty cycles. When Vc reaches or exceeds Vcq, the curve 702 shows thecase for continued operation at a gain value of 1.0, and curves 703, 704and 705 respectively show the selective linearization operation at gainvalues G_(A) of 2.0, 4.0 and 5.0.

The selective conversion gain linearization shown in the curves 703-705provides advantages in using the linearization apparatus 110 withrespect to control stability and reduced EMI, noise and jitter relativeto the unmodified operation shown in curve 702 for duty cycles above thethreshold Vcq. In other examples, the linearization apparatus 110 may beemployed in combination with other DC-DC converter stage circuitryoperable above unity conversion gain, for example Cuk converters (notshown).

FIG. 8 illustrates a DC-DC power conversion system example 800 with alinearization apparatus 110 providing the functionality described above,where the system in this case uses digital control. The power stage 101(e.g., a boost converter, a flyback converter, a buck-boost converter,Cuk converter) receives an input voltage signal VIN from a source 102and provides an output voltage signal VOUT to a load (not shown). Afeedback circuit including a resistive voltage divider formed by R2 andR3 provides a feedback voltage signal VFB to an analog-to-digital (A/D)converter 802. The A/D converter 802 provides a series of feedbackvoltage samples VFB[n] to a digital compensator circuit 804 operatingaccording to a first clock input signal CLK1 to provide compensatedvoltage sample values VC[n] to a digital linearization circuit 110operated according to a second input clock signal CLK2. Thelinearization circuit 110 in one example is adjustable or programmable,and receives a threshold setting value VcqS 810 from an electronicmemory 806 and/or a gain value setting G_(A)S 812 from the memory 806.

In operation, the linearization circuit 110 provides a PWM signal 122 toa driver and dead-time logic circuit 104 to provide pulse widthmodulated switching control signals d and d′ along lines 106 and 108 tooperate the power stage 101 as described above. In addition, thedigitally implemented linearization circuit 110 is programmed orotherwise configured to provide the PWM signal 122 according to theoriginal or digitally compensated values VFB[n] or VC[n] for a firstrange of compensation values. After a threshold compensation value hasbeen reached or exceeded, the linearization circuit 110 linearizes thesystem conversion gain M by modifying the values VFB[n] or VC[n]according to the VcqS and G_(A)S settings 810 and 812 from the memory806.

FIG. 9 shows an example adjustable threshold circuit 140 allowingadjustment of the threshold signal Vcq in the systems 100 of FIGS. 1, 2and 6. In this example, a reference voltage source 902 provides athreshold reference voltage VREF2. The circuit 140 includes a switchableresistive voltage divider circuit with an integer number N thresholdadjustment switches STH0 through STHN-1 connected in a binary weightedfashion with a series connection of threshold voltage divider top sideresistances RTHT. The top resistance and a bottom side voltage dividerresistance RTHB provide the threshold signal Vcq at a select fraction ofthe threshold reference voltage VREF2. Any resistor sizes and relativeresistances can be used. In one example, the bottom resistance RTHB hasa value equal to N times the value of the individual top sideresistances TRTH (RTHB=N*RTHT). In this example, the threshold voltagesignal Vcq as a value equal to VREF2 times the total top side resistancefor the top side resistors RTHT not bypassed or shorted by acorresponding one of the switches STH, divided by the sum of the bottomside resistance and the non-shorted top side resistance, allowingtailoring by activation of certain switches STH0 through STHN-1. In thisexample, a threshold select circuit 904 receives a desired thresholdselection signal 810 (VcqS) from the memory 806 and provides a set of Ncorresponding binary-weighted threshold control signals CTH0, CTH1 . . .CTHN-1 to selectively tailor the adjustable threshold signal amplitudeVcq.

FIG. 10 illustrates an adjustable gain circuit example 150 forselectively adjusting the gain value G_(A) in the system 100 of FIGS. 1,2 and 6. In this case, the top and bottom side current mirrors includeat least one switch-selectable branch with a mirrored pair of P-channeland N-channel transistors MP4 and MN2 and corresponding plus and minusside switches for selectively modifying the current I1 provided to thegain circuit output node 152. A gain select circuit 1004 in one examplereceives a gain value G_(A)S select signal 812 (G_(A)S) from the memory806 and generates a corresponding set of K gain control signals CG0, GC1. . . CGK-1 to operate the corresponding pairs of branch switches SGPand SGM. In one example, MP4-0, SGP0, SGM0 and MN2-0 form a firstswitchable branch operated according to control signal CG0. Anotherselectable branch circuit includes MP4-1, SGP1, SGM1 and MN2-1controlled by a signal CG1. An integer number K switch selectable branchcircuits are provided including a final branch formed by MP4-K-1,SGPK-1, SGMK-1 and MN2-K-1 and controlled by signal CGK-1, where K is aninteger greater than 0. In this example, the selectable positive sidetransistors MP4-0, MP4-1 . . . MP4-K-1 are sized the same as MP4 (N-K)and the negative side mirror transistors MN2-0, MN2-1 . . . MN2-K-1 aresized the same as MN2 (N-K), with the other mirror transistors MP1-MP3and MN1 sized the same as one another (N times the unit w/l). The gainselect circuit 1004 in this example provides a range of adjustable orprogrammable gain values G_(A) ranging from a value of 1.0 with all theswitches closed, a gain of 2.0 with only one pair of switches open, again of 4.0 with two pairs of switches open, etc.

FIG. 11 illustrates a method 1100 of linearizing a conversion gain of aDC-DC converter circuit (e.g., circuits 101, 601 above). At 1102, afirst compensation signal is received, such as a voltage signal Vcdescribed above representing a difference between a DC-DC converteroutput signal (e.g., VOUT in FIGS. 1 and 2) and a reference signal(e.g., VREF). A PWM signal is generated at 1104 according to a secondcompensation signal (Vca), and the DC-DC converter is operated at 1106according to the PWM signal. In this manner, one or more switches of aDC-DC converter are operated at least partially according to the secondcompensation signal Vca. At 1108 in FIG. 11, a gain circuit outputsignal is generated (e.g., I1) according to a non-zero gain value (e.g.,G_(A)) and the difference between the first compensation signal and athreshold signal (I1 is proportional to (Vc−Vcg)/G_(A)). At 1110, adetermination is made as to whether Vc is less than the threshold signalVcq. If so (YES at 1110), the second compensation signal Vca isgenerated at 1112 based on Vc, and the process 1100 returns to 1102.Otherwise (NO at 1110), Vca is generated at 1114 based on the thresholdsignal Vcq and the gain circuit output signal I1 to selectivelylinearize the conversion gain of the DC-DC converter.

The present disclosure provides systems as well as apparatus and methodswhich can be employed to linearize the conversion gain duty cycle abovea chosen operating point (D_(Q)). The disclosed examples work well inclosed loop systems unlike most other solutions, and eliminate or reducethe problems associated high conversion ratio operation. Disclosedexamples also allow the flexibility to choose or parameterize thethreshold and gain values D_(Q) and G_(A). The linearization apparatus110 in certain examples facilitates extension of the range of operationof the converter circuit 101, 601 with a slope function of chosen gain,where certain implementations allow choice of the threshold signal andthe gain value D_(Q) (Vcq) and G_(A) to implement any desired M v/s Dcurve. The resulting linearized conversion gain profile M facilitatesreliable, stable high boost ratio in a single converter stage. In otherexamples, multiple thresholds and corresponding gain values can beimplemented in a single linearization apparatus 110.

The disclosed solution facilitates desensitized operation of theconverter system 100 with respect to transient line/load perturbationsduring high duty cycle operation due to the proposed linearized approachin closed loop operation. In addition, the concepts of the presentdisclosure may also be applied in open loop implementations. Also, thedisclosed adjustable gain and/or threshold concepts may be implementedfor on-chip programmability to facilitate use in various applications,such as driving LED strings of different lengths in lightingapplications. In addition, the disclosed concepts and apparatus may beemployed in boost as well as other DC-DC converter configurations, suchas flyback, non-inverting buck-boost and Cuk designs operating in aboost mode. Also, the linearization apparatus 110 and the disclosedconcepts can be used in existing open loop or closed loop designs inboth continuous conduction mode (CCM) or discontinuous conduction mode(DCM) power conversion systems.

The above examples are merely illustrative of several possibleembodiments of various aspects of the present disclosure, whereinequivalent alterations and/or modifications will occur to others skilledin the art upon reading and understanding this specification and theannexed drawings. Modifications are possible in the describedembodiments, and other embodiments are possible, within the scope of theclaims. In addition, although a particular feature of the disclosure mayhave been disclosed with respect to only one of multipleimplementations, such feature may be combined with one or more otherfeatures of other embodiments as may be desired and advantageous for anygiven or particular application. Also, to the extent that the terms“including”, “includes”, “having”, “has”, “with”, or variants thereofare used in the detailed description and/or in the claims, such termsare intended to be inclusive in a manner similar to the term“comprising”.

What is claimed is:
 1. A circuit, comprising: a comparator configured togenerate an adjusted compensation signal when a first compensationvoltage is greater than a linearization threshold voltage; and aswitching circuit coupled to the comparator, and configured to generatea second compensation voltage responsive to the adjusted compensationsignal, the second compensation voltage aggregating the firstcompensation voltage with a scaled difference between the firstcompensation voltage and the linearization threshold voltage.
 2. Thecircuit of claim 1, wherein the first compensation voltage correspondsto an error between a reference voltage and a feedback voltage receivedfrom an output of a power converter.
 3. The circuit of claim 1, whereinthe linearization threshold voltage corresponds to a duty cycle of aswitch mode power converter, and the duty cycle is equal to or greaterthan 0.5.
 4. The circuit of claim 1, further comprising: a gain circuitconfigured to convert the first compensation voltage and thelinearization threshold voltage to a first current and a second currentrespectively, and generate an output current scaling a differentialcurrent between the first and second currents by a non-zero gain value,wherein the switching circuit configured to generate a scaleddifferential voltage based on the output current and a resistor andaggregate the scaled differential voltage to the first compensationvoltage in generating the second compensation voltage.
 5. The circuit ofclaim 4, wherein the gain circuit includes a differential current mirrorcircuit having: a first current path configured to conduct the firstcurrent; a second current path configured to conduct the second current;and a third current path configured to scale the differential currentbetween the first and second currents by the non-zero gain value.
 6. Thecircuit of claim 1, further comprising: a gain circuit having: atransconductance amplifier having first and second outputs to deliverfirst and second currents converted from the first compensation voltageand the linearization threshold voltage respective respectively; a firstPMOS transistor having a first drain coupled to the second output, and afirst gate coupled to the first drain; a second PMOS transistor having asecond drain, and a second gate coupled to the first gate; a third PMOStransistor having a third drain coupled to the first output, and a thirdgate coupled to the third drain; a fourth PMOS transistor having afourth drain coupled, and a fourth gate coupled to the third gate; afirst NMOS transistor having a fifth drain coupled to the second drain,and a fifth gate coupled to the fifth drain; and a second NMOStransistor having a sixth drain coupled to the fourth drain, and a sixthgate coupled to the fifth gate; and a current output terminal coupled tothe fourth drain and the sixth drain.
 7. The circuit of claim 1, whereinthe switching circuit includes: a resistor; a first switch configured topass a first current in response to the adjusted compensation signal,the first current proportional to the scaled difference between thefirst compensation voltage and the linearization threshold voltage, andthe first current conducted via the resistor; and a second switchconfigured to pass a second current in response to the adjustedcompensation signal, the second current proportional to thelinearization threshold voltage and conducted via the resistor.
 8. Thecircuit of claim 1, wherein: the comparator is configured to generate aunadjusted compensation signal when the first compensation voltage isless than the linearization threshold voltage; and the switching circuitincludes: a resistor; a switch configured to pass a current in responseto the unadjusted compensation signal, the current proportional to thefirst compensation voltage and conducted via the resistor.
 9. A circuit,comprising: a comparator configured to assert a first mode signal when afirst compensation voltage is less than a linearization thresholdvoltage, and assert a second mode signal when the first compensationvoltage is greater than the linearization threshold voltage; and aswitching circuit coupled to the comparator, and configured to: selectthe first compensation voltage when the first mode signal is asserted;and select a second compensation voltage when the second mode signal isasserted, the second compensation voltage aggregating the firstcompensation voltage with a scaled difference between the firstcompensation voltage and the linearization threshold voltage.
 10. Thecircuit of claim 9, wherein the first compensation voltage correspondsto an error between a reference voltage and a feedback voltage receivedfrom an output of a power converter.
 11. The circuit of claim 9, whereinthe linearization threshold voltage corresponds to a duty cycle of aswitch mode power converter, and the duty cycle is equal to or greaterthan 0.5.
 12. The circuit of claim 9, wherein the switching circuitincludes: a resistor; a first switch configured to pass a first currentwhen the second mode signal is asserted, the first current proportionalto the scaled difference between the first compensation voltage and thelinearization threshold voltage, and the first current conducted via theresistor; a second switch configured to pass a second current when thefirst mode signal is asserted, the second current proportional to thefirst compensation voltage and conducted via the resistor; and a thirdswitch configured to pass a third current when the second mode signal isasserted, the third current proportional to the linearization thresholdvoltage and conducted via the resistor.
 13. The circuit of claim 12,further comprising: a ramp generator circuit; and a pulse widthmodulation (PWM) generator having a non-inverting input coupled to theresistor, an inverting input coupled to the ramp generator circuit, andan output configured to deliver a PWM signal based on a comparisonbetween the non-inverting input and the inverting input.
 14. The circuitof claim 9, further comprising: a gain circuit configured to convert thefirst compensation voltage and the linearization threshold voltage to afirst current and a second current respectively, and generate an outputcurrent scaling a differential current between the first and secondcurrents by a non-zero gain value, wherein the switching circuitconfigured to generate a scaled differential voltage based on the outputcurrent and a resistor and aggregate the scaled differential voltage tothe first compensation voltage in generating the second compensationvoltage.
 15. The circuit of claim 14, wherein the gain circuit includesa differential current mirror circuit having: a first current pathconfigured to conduct the first current; a second current pathconfigured to conduct the second current; and a third current pathconfigured to scale the differential current between the first andsecond currents by the non-zero gain value.
 16. A circuit, comprising: acomparator configured to assert a first mode signal when a firstcompensation voltage is less than a linearization threshold voltage, andassert a second mode signal when the first compensation voltage isgreater than the linearization threshold voltage; and a gain circuitconfigured to convert the first compensation voltage and thelinearization threshold voltage to a first current and a second currentrespectively, and generate an output current scaling a differentialcurrent between the first and second currents by a non-zero gain value;and a switching circuit coupled to the comparator, and configured to:select the first compensation voltage when the first mode signal isasserted; and select a second compensation voltage when the second modesignal is asserted, the second compensation voltage summing the firstcompensation voltage with a scaled difference voltage based on theoutput current.
 17. The circuit of claim 16, wherein the firstcompensation voltage corresponds to an error between a reference voltageand a feedback voltage received from an output of a power converter. 18.The circuit of claim 16, wherein the linearization threshold voltagecorresponds to a duty cycle of a switch mode power converter, and theduty cycle is equal to or greater than 0.5.
 19. The circuit of claim 16,wherein the gain circuit includes a differential current mirror circuithaving: a first current path configured to conduct the first current; asecond current path configured to conduct the second current; and athird current path configured to scale the differential current betweenthe first and second currents by the non-zero gain value.
 20. Thecircuit of claim 16, wherein the gain circuit includes: atransconductance amplifier having first and second outputs to deliverfirst and second currents converted from the first compensation voltageand the linearization threshold voltage respective respectively; a firstPMOS transistor having a first drain coupled to the second output, and afirst gate coupled to the first drain; a second PMOS transistor having asecond drain, and a second gate coupled to the first gate; a third PMOStransistor having a third drain coupled to the first output, and a thirdgate coupled to the third drain; a fourth PMOS transistor having afourth drain coupled, and a fourth gate coupled to the third gate; afirst NMOS transistor having a fifth drain coupled to the second drain,and a fifth gate coupled to the fifth drain; and a second NMOStransistor having a sixth drain coupled to the fourth drain, and a sixthgate coupled to the fifth gate; and a current output terminal coupled tothe fourth drain and the sixth drain.